Freescale Semiconductor, Inc., 2006. All rights reserved.Freescale Semiconductor [625149]

© Freescale Semiconductor, Inc., 2006. All rights reserved.Freescale Semiconductor
Application NoteAN3234
Rev. 0, 02/2006
Table of Contents The latest trend in washing machine design is to replace
traditional drive systems wi th modern, electronically
controlled, brushless driv es. In the past, washing
machine designs employed two widely used drive
systems. The older designs use electromechanically
controlled two-speed single phase AC induction motors.
This kind of drive system is no longer used for new
machines and is only found in the least expensive washer
models. The majority of wash ers have universal brushed
motors with Triode Alternating Current switch (TRIAC)
control. However, with the advent of new electronic
devices, these drives are becoming out-of-date. A new
generation of washing machines will be designed with
brushless three-phase motors. The best candidat: [anonimizat], and this requir es microcontroller based
solutions. DSP-based devices are preferred because of
the real-time signal processi ng demands from AC motor
control applications. This a pplication note presents the
AC induction motor alternative, focusing on the
description of suitable control algorithms and its
implementation in a real washer application.1 Drive Features . . . . . . . . . . . . . . . . . . . . . . . . . . . 2
2 Indirect Vector Control . . . . . . . . . . . . . . . . . . . . . 33 Control Algorithm Overview . . . . . . . . . . . . . . . . . 5
3.1 Motor Model Block . . . . . . . . . . . . . . . . . . . . . 6
3.2 Rotor Flux Model . . . . . . . . . . . . . . . . . . . . . . 73.3 Space Vector Modulation. . . . . . . . . . . . . . . . 7
3.4 Current Control Loop . . . . . . . . . . . . . . . . . . . 9
3.5 Torque Producing Component Estimation
Block . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
3.6 Stator Loss Optimization Block . . . . . . . . . . 12
3.7 Field-Weakening Control Block . . . . . . . . . . 143.8 Speed Control Loop. . . . . . . . . . . . . . . . . . . 15
3.9 Quadrature Component Evaluation Block . . 16
4 Washing Machine Drive Operating Modes . . . . . 165 User Control Interface. . . . . . . . . . . . . . . . . . . . . 18
6 Washer Drive Parameters Tuning. . . . . . . . . . . . 18
7 Freescale Semiconductor Support . . . . . . . . . . . 198 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
9 Glossary of Symbols . . . . . . . . . . . . . . . . . . . . . . 20Washing Machine Three-Phase
AC Induction Motor Drive
Based on MC56F8013
by: Petr Stekl
Freescale Semiconductor, Inc.

Washing Machine Three-Phase AC Induction Motor Drive, Rev. 0Drive Features
Freescale Semiconductor 21 Drive Features
The three-phase AC induction washing machine driv e responds to the new market demands for higher
performing appliances. The aim is to provide maximum drive performance at a competitive price, served
particularly well by Frees cale Semiconductor's recently introduc ed 56F801x family of hybrid digital
signal processor/microcontroller (DSP/MCU) embedded controllers. An example drive design based on
the MC56F8013 offers the product designer plenty of co mputing power with adva nced peripherals at a
very good price/performance ra tio. The most important feat ures of the drive include:
• Three-phase AC induction motor• Cost-efficient tachogenerator on motor shaft for speed sensing
• Indirect vector control algorithm• Speed range 0 – 20000 RPM (motor speed), 0 – 2000 RPM (drum speed)• Reconstruction of three-phase curr ents from DC-bus shunt resistor
• Non-recuperative braking and deceleration control
• Loss-minimizing control• Over-current, over-voltage and under-voltage protection• Out-of-balance detection for spin dry• Serial RS232 control interface
It should be highlighted, the presented drive was deve loped with considerable unique requirements of the
washing machine application. The drive is designed to run a very wide range of speeds, from 0 – 20000
RPM. It is optimized to accept a wide range of loads. This feature refl ects the condition of a real washer,
required to run reliably with both an empty drum a nd a drum fully stacked with wet and heavy clothes.
Another specific feature of the wash er application is the ability to de velop a high start-up torque for the
motor to force the full drum to move. As the efficien cy of washing depends on pr ecise speed control of the
washer drum, the presented drive co mes with a PID speed co ntrol closed loop. Thanks to the inner closed
current control loop, the presented dr ive features high dynamics to achieve top performance control. It is
required to shorten the washing cycle as much as possible. A shortened washing cycle is achieved by using
a non-recuperative braking algorithm to stop the drum when it finishes a high speed spin-dry; a very
important aspect is energy efficiency. The presented drive comes with a loss-minimizing algorithm to run
at an optimum operating point and so save on valued energy. Thanks to selected control techniques, the
drive shows high immunity to motor parameter tole rances and to changes during its operation and
life-time.
Emphasis was put on the design of a product capable of competing in a market as cost sensitive as white
goods market require. Consider ing cost effectiveness, the drive reduc es the number of current sensors. The
number of current sensors for sensing the motor current is re duced from three to a single-shunt resistor on
the DC-Bus. The three-phase motor currents are recons tructed from the DC-Bus cu rrent using an advanced
reconstruction technique.

Indirect Vector Control
Washing Machine Three-Phase AC Induction Motor Drive, Rev. 0
Freescale Semiconductor 32 Indirect Vector Control
Indirect vector control belongs to the family of vector control techni ques. Compared to direct vector
control structures, indirect vector control does not require direct real -time calculation of rotor flux from
motor currents and voltages. Due to this indirect vect or control, it is not possi ble to obtain instantaneous
values of the rotor flux space vector components. It is, however, still possible to control motor excitation
and torque independently. In a steady state we will achieve the same pe rformance as with direct vector
control. In a transient state, a cert ain error can be observed when compared to direct control. This error,
however, for most applications is neglig ible, including the washing machine drive.
The control technique algorithm was developed consider ing an equivalent steady state circuit, shown in
Figure 1 .
Figure 1. Induction Motor Equivalent Circuit
The equivalent circuit is valid in the steady state only. A full description of the induction motor model
gives a set of motor equations ( Equation 1 – Equation 9 ) expressed in a rotational d,q-reference frame.
Eqn. 1
Eqn. 2Eqn. 3Eqn. 4
Eqn. 5
Eqn. 6Eqn. 7Eqn. 8
Eqn. 9Rs
Rr.(1-s)/sLsσ Lrσ
LmRr
usd RsisdtddΨsdωsΨsq – + =
usq RsisqtddΨsqωsΨsd – + =
urd 0 RrirdtddΨrd ωsω–() Ψrq – + ==
urq 0 RrirqtddΨrq ωsω–() Ψrd ++ ==
Ψsd LsisdLmird + =
Ψsq LsisqLmirq + =
Ψrd LrirdLmisd + =
Ψrq LrirqLmisq + =
T3
2–ppΨrdisq =

Washing Machine Three-Phase AC Induction Motor Drive, Rev. 0Indirect Vector Control
Freescale Semiconductor 4If you are looking for more theory on the field oriented control of a three-phase ac induction motor, please
refer to [ 1]. For a glossary of the symbols used, please, refer to Section 9 .
The assumption in the indirect vector control algorithm is the rotor flux space-vector size and position are
defined by the applied motor voltage and current. Based on the induction moto r model, we are able to draw
a space-vector diagram; see Figure 2 . As can be seen in the space-vector diagram, the position and size of
the rotor flux is fully determined by the voltage and cu rrent vectors for the given motor. Indirect vector
control uses this fact to control th e space-vector quantities of the motor.
The indirect vector control algorithm for an induction motor impl emented in the presen ted design is based
on the following assumptions:
• the instantaneous stator voltage vector amplitude is calculated with high accuracy corresponding
to the actual motor operating point
• having a precise stator volta ge generated on the motor, a good estimation of the motor
magnetizing flux is achieved
• the stator current of the motor is set by the PI controller to maintain the required value given by
the quadrature and direct components
• if points 1-3 are satisfied, a dir ect-axis component of the stator current is obtained, the same as
required by the control
• if the stator current amplitude Is and direct-axis component Isd are kept at the required values, the
quadrature-axis component of the stator current is Isq
The above mentioned principles of the control technique can be unders tood easily with the help of the
induction motor vect or diagram in Figure 2 . The diagram displays the relatio ns between the stator voltage
(Us), stator current (Is), and the rotor, stator and magnetizing flux ( Ψr,Ψs,Ψm).
Figure 2. Induction Motor Space-Vector Diagramψmψsψsσ
ψrψrσd-axisq-axis β-axis
α-axisωre
θre
flux-controllingtorque-controlling
Us
IrIsq
IsdImIs

Control Algori thm Overview
Washing Machine Three-Phase AC Induction Motor Drive, Rev. 0
Freescale Semiconductor 53 Control Algorithm Overview
Tough requirements placed on a washing machine driv e call for a high-performance control algorithm.
Good candidates for this job are vector control t echniques. The presented algorithm is based on the
implementation of the indirect vect or control technique. The control st ructure overview is illustrated in
Figure 3 . Similarly, as with othe r vector control oriented techniques, the implemented algorithm is able to
control the excitation and torque of the induction motor separately. The idea of indirect vector control is
based on the indirect contro l of motor flux through the control of mo tor voltage and current. The torque
command for the control algorit hm is taken from the PI D speed controller. The re ference for motor flux is
set by the Loss Optimization block for speeds belo w the nominal. For speeds and voltages above the
nominal, the Field-Weakening block ta kes over the setting of th e flux reference. The aim of the control is
the regulation of the motor (washer drum) speed. The speed command value is set by high level control,
i.e. the washing programmer.
To achieve the goal of the induction motor control, the algorithm utili zes a set of feedback signals. The
essential feedback signals are as fo llows: DC-bus voltage, three-phase stator current reconstructed from
DC-bus current, motor speed. For correct operation, the presented control structure requires a speed sensor
on the motor shaft. For this purpose a tachogenerator is used.
Figure 3. Control Algorithm OverviewMeasured Speedf_motor+Flux ModelFlux Model
Motor Model
SVM
f_stator
current_statorIsq_estimIsdwIsdw
-+speed_cmd
PIDvoltage_stator
PI
–
3ph current
reconstructionDC-Bus CurrentFeedFwdTorque
to IsqwIsdmax
Isq Estim.Field-Weakening
Control
PWM
f_stator
voltage_stator
IsdmaxTorqueLoss
Optimisationf_slip
2 2Isdw Isqw +
+ ++
Isqw-+
Speed ControllerCurrent ControllerΨrw
Ψrw

Washing Machine Three-Phase AC Induction Motor Drive, Rev. 0Control Algorithm Overview
Freescale Semiconductor 63.1 Motor Model Block
The block diagram of the control algorithm ( Figure 3 ) illustrates the stator vol tage amplitude evaluated by
the Motor Model block. The precise stator voltage amplitude is cal culated based on the motor model
equations ( Equation 10 , Equation 11 , Equation 12 ) from the required quadratur e and direct components of
the stator current, required rotor flux, ac tual motor slip and stator frequency.
Figure 4. Motor Model Block
Eqn. 10
Eqn. 11
Eqn. 12
The above equations are derived from the stator and rotor equations of the induction motor, expressed in
a synchronous rotational reference frame (d,q). Assu mptions considered for the Motor Model block
equations are as follows:
Eqn. 13
Eqn. 14
Eqn. 15
Eqn. 16
The stator voltage amplitude evalua tion is one of the key assumptions in indirect vector control. The
advantage of the above equations is a low sensitivity to ch anges in rotor resistance Rr. The rotor resistance
is highly temperature depende nt and can vary considerab ly during the motor working cycle. The effect of
this change on the overall voltage space-vector amplitude is, however, ne gligible.The influence of stator Motor Model
f_statorΨrw
voltage_stator
f_slipIsqwIsdw
Usd RsIsdωsLsσIsq – ωsLrσωr
Rr––Ψr – =
Usq RsIsqωsLsσIsdωsΨr ++ =
Us Usd2Usd2+ =
Ψrq 0=
d
dt––Ψr 0=
ωs2πfsωsΨ≅ =
ωrωsω– ωrΨωsΨω– = ≅ =

Control Algori thm Overview
Washing Machine Three-Phase AC Induction Motor Drive, Rev. 0
Freescale Semiconductor 7resistance Rs is also acceptable at non-zero speeds due to much higher component of BEMF voltage. At
zero speeds negative influence of stator resistance change can be easily compensated at startup.
3.2 Rotor Flux Model
The Rotor Flux Model block defines the relation between the direct-axis component of the stator current
and the rotor magnetizing flux vector amplitude. The rotor flux model resp ects the rotor flux time-constant
as well as the non-linear ma gnetizing curve of the real induction mach ine. The block solves the differential
equation ( Equation 15 ).
Figure 5. Rotor Flux Model Block
The block has one input variable: the direct -axis component of the stator current (Isd) and one output
variable: the actual value of the rotor-flux vector ( Ψrw); see Figure 5 .
Eqn. 17
The algorithm of solving the Equation 15 is graphically depicted in Figure 6 .
Figure 6. Rotor Flux Algorithm
3.3 Space Vector Modulation
A motor voltage evaluated in the Motor Model Block is generated by the three-phase voltage source
inverter and applied to the stator of the motor. The voltage source inverter converts the DC-Bus voltage to Flux ModelFlux ModelΨrw Isdw
Ψr R∫rIsdmagCurveInv Ψr() –() dt ⋅ =
Rr ∫+
–
magCurveInvIsd Ψrw

Washing Machine Three-Phase AC Induction Motor Drive, Rev. 0Control Algorithm Overview
Freescale Semiconductor 8AC voltage of the required amplitude and frequency. Typical three-phase inverter topology is illustrated
in Figure 7 .
Figure 7. Three-Phase Voltage Source Inverter
Space Vector Modulation (SVM) can directly transform the stator voltage vectors from α,β-coordinate
system to pulse width m odulation (PWM) signals (duty cycle valu es). It can generate eight possible
switching states (vectors) with a three-phase voltage source inverter. They are given by combinations of
the corresponding power switches.A gr aphical representation of all comb inations is the hexagon shown in
Figure 8 . There are six non- zero vectors, U0, U60, U120, U180, U240, U300, and two zero vectors, O000 and
O111, defined in α,β coordinates.
Figure 8. Basic Space Vectors and Voltage Vector Projection
T
1
T
2T
3
T
4T
5
T
6
3-Phase
AC MotorPh. A Ph. CPh. BC+ DC-Bus
– DC-Bus+
uβ
uαMaximal phase
voltage magnitude = 1Basic Space Vector
α-axisβ-axis
II.
III.
IV.
V.VI.U
(110)60U
(010)120
U
(011)180
O
(111)000 O
(000) 111
U
(001)240 U
(101)300U
(100)0
[2/ 3,0]√ [-2/ 3,0]√[1/ 3,1]√
[-1/ 3,1]√[1/ 3,-1]√
[-1/ 3,-1]√T/ T *0U0T/ T * U60 60
US
Voltage vector components
in α, β axis30 degrees

Control Algori thm Overview
Washing Machine Three-Phase AC Induction Motor Drive, Rev. 0
Freescale Semiconductor 9SVM is a technique used as a direct bridge between vector control ( voltage space vect or) and PWM. The
SVM technique consists of three steps:
• sector identification• space voltage vector decom position into dir ections of sector base vectors U
x, Ux±60
• PWM duty cycle calculation
The principle of SVM is the appl ication of the voltage vectors UXXX and OXXX for certain instances in
such a way the “mean vector” of the PWM period TPWM is equal to the desired voltage vector. The
implemented SVM technique fully util izes the DC-Bus voltage for generation of the output stator voltage.
The maximum amplitude of the output phase voltage is . For more
information on space vector modulation technique, please, refer to [ 1].
The DC-Bus voltage level is not cons tant. Its level can vary with differ ent power line conditions. Also, if
the DC-Bus is supplied from a rec tified single-phase AC supply, the DC -Bus voltage contains a voltage
ripple, potentially several te ns of volts. The DC-Bus voltage ripples can create distortion to the generated
sinusoidal output. Therefore, a “DC-Bus Ripple Eli mination” algorithm is implemented, removing the
distortion from the output voltage . The algorithm is shown in Figure 9 .
Figure 9. DC-Bus Ripple Elimination
Firstly, the required output voltage is divided by a fi ltered value of half the actual DC-Bus voltage. The
result is multiplied by the invers e value of the modulation index to get the corresponding duty-cycle.
Finally, the Space Vector Modulation algorithm evalua tes all six PWM registers to generate the required
voltage vector at the output of the three-phase inverter bridge.
3.4 Current Control Loop
The indirect vector control algorit hm requires a stator cu rrent control loop, discu ssed earlier. The current
control loop block diagram is illustrated in Figure 10 . Control of the stator current is achieved with a
PI-controller. The controller output se ts the rotor slip frequency to c ontrol the stator current amplitude.
From the induction motor theory it is possible to sh ow the motor slip frequency is proportional to the
torque-producing component of the stator current Isq. Hence, the current control loop provides only a Uphmax _amplitude2
3––-UDC Bus– 2⁄ ()⋅ =
LP Filter
Ustator÷UDC-Bus
SVM
2321
3
2––-

Washing Machine Three-Phase AC Induction Motor Drive, Rev. 0Control Algorithm Overview
Freescale Semiconductor 10control of the quadrature axis compone nt of the stator current. The feed back quantity of the PI controller
is, anyway, an amplitude (size) of th e space vector of the stator current Is. The direct-axis (flux-producing)
component Isd of the stator current is controll ed via the change in the stator voltage applied to the motor,
evaluated in the Motor Model Block ( Section 3.1 ).
The block diagram in Figure 10 illustrates the controller algorithm takes the required direct-axis ( Isdw) and
required quadrature axis ( Isqw) components of the stator current as reference values. Having the current
components, the amplitude ( Isw) of the required stator current space vector is evaluated according to the
equation:
Eqn. 18
The amplitude of the re quired stator current ( Isw) is compared to the feedback of the actual stator current
amplitude sensed on the motor ( Is).
Figure 10. Current Controller Loop
To reduce the number of current sensor s, the three-phase stator currents are measured by means of a single
DC-Bus current shunt sensor (see Figure 11 ). The DC-Bus current pulses are sampled at exactly timed
intervals. Based on the actual comb ination of switches, the three-phase currents of the stator are
reconstructed. The three-phase current s are transformed into alpha, beta components of the space-vector
in the stationary reference frame. Having the alpha, be ta components, the actual vector size (amplitude) is
evaluated and used as a feedback signal for the PI controller.Isw Isdw Isqw+ =
Measured Speed+f_stator
Stator Current
VectorPI
-2 2Isdw Isqw +
FeedFwd
3ph current
reconstructionIsdw
Isqw
DC-Bus Current
Forward
Clarke+-+
++f_slip
ΨrwIsIs

Control Algori thm Overview
Washing Machine Three-Phase AC Induction Motor Drive, Rev. 0
Freescale Semiconductor 11Figure 11. Single Shunt Current Sensing Approach
To improve dynamic beha vior of the current contro l loop, there is an additiona l feed forward implemented.
The instantaneous rotor slip is eval uated from the required quadrature axis component of the stator current
(Isqw). The feed forward algorithm is evaluated according to:
Eqn. 19
The constant KM is a motor dependent constant. It can be eval uated from the particular motor parameters.
The constant KM can be evaluated according to this formula:
Eqn. 20
Because the estimated motor parameters can be slightly different from the real ones, the application can
be run in a calibration mode to ac hieve the best approximation of KM.
3.5 Torque Producing Component Estimation Block
The indirect field oriented contro l algorithm does not evaluate the stator current components in the
rotational reference frame (d,q). Ther efore, the position of the space ve ctor of the rotor magnetizing flux
is not required to be evaluated. This brings the advantage of lower demands on the microcontroller
computational resources. Als o, because of high sensitivit y of the rotor flux model to motor parameters, it
makes the control of the motor torque algorithm le ss dependent on swinging motor parameters. Because
the quadrature axis component of the stator current ( Isq) is one of the input quantities for stator voltage
evaluation, it is necessary to estima te this quantity with help of known quantities. For estimation, use the
same dependency between roto r the slip frequency and th e quadrature axis component of the stator current,
as in the case of feed forward ( Equation 19 ). The formula can be trasnformed in the following way:
Eqn. 21UdcUdc
fslip KMIsqw
Ψrw–––⋅ =
KMLmRr⋅
2πL⋅r–––––-=
Isq_estimatedΨrwfslip⋅
KM–––––––- – =

Washing Machine Three-Phase AC Induction Motor Drive, Rev. 0Control Algorithm Overview
Freescale Semiconductor 12The diagram of the quadrature axis comp onent estimation block is illustrated in Figure 12 . It has two input
variables:
• actual value of the rotor slip frequency (fslip)
• required size (amplitude) of the rotor magnetizing flux space vector
Figure 12. Torque producing Estimation Block
Output from the quadrature axis com ponent estimation block is taken as an input to the Motor Model Block
(see Figure 3 ).
3.6 Stator Loss Optimization Block
The aim of the Stator Lo ss Optimization Block (see Figure 13 ) is to minimize power losses in the induction
motor. Power losses in a motor are de fined by the stator curren t flowing to the motor. For every motor load
it is possible to find an optimal point, at which the losses are minimal and the motor is operating with
highest efficiency. The stator cu rrent of the induction motor has two components – torque producing
component and a magnetizing flux pr oducing component. The motor tor que is a proportional product of
those components as indicat ed by following formula:
Eqn. 22
Common field-oriented control techni ques keep the motor flux constant at its nominal value. The required
torque is then set by only controlli ng of the torque produci ng component. In conditions of a low load this
approach is not efficient. A motor flux maintained at its nominal value generate s additional losses in the
stator windings.
Figure 13. Stator Winding Loss Optimization Block
The aim of the Stator Loss Optimization algorithm is to control the motor magnetizing current to reduce
stator losses. The motor magnetizing flux is not mainta ined as a constant. The algorithm finds the most
optimal operating point for the actual motor load. This helps to improve the overall efficiency of the motor.
The copper losses in the stator windings ca n be evaluated according to the formula:Isq
EstIsq_estimated
f_slipΨrw
TMotor IsdIsq⋅ ≈
Loss
OptimisationLoss
OptimisationIsdmax Torque

Control Algori thm Overview
Washing Machine Three-Phase AC Induction Motor Drive, Rev. 0
Freescale Semiconductor 13Eqn. 23
Considering Equation 9 , Equation 7 and Equation 23 a condition of minimum losse s in the stator windings
is achieved.
Eqn. 24
For the motor torque, the following formula can be evaluated based on Equation 9 :
Eqn. 25
For the steady state it holds:
Eqn. 26
Combining Equation 24 , Equation 25 and Equation 26 a formula results, used to evaluate the optimal
magnetizing flux for a given motor torque:
Eqn. 27
Equation 27 is evaluated within the Stator Loss Optimiza tion Block. It sets the optimal magnetization flux
level for the actual motor torque to minimize losses in the stator windings. The internal structure of the
block is illustrated in Figure 14 .
Figure 14. Internal Structure of Stator Loss Optimization Block
The algorithm in Figure 14 first evaluates the optimal level of the magnetizing flux Ψr_optimal . From the
magnetization curve, the corresponding flux produc ing component of the stator current Isdmax is
determined. This value is taken as a reference defining the maximum magnetiz ing current meeting the
optimal operating point of the induction motor. It se rves as an input to the Field-Weakening Block.
This type of stator windings loss opt imization is often called “Optimal Slip Control.” This is because if
condition Equation 24 is fulfilled, the motor slip frequency is ke pt constant at all operating points (motor
speed and load). The optimal slip can be evalua ted for the given motor according to the formula:
Eqn. 28Pstator RsIsd2Isq2+()≈ ∆
Isd
Isq––1=
T3
2–ppLm2
Lr––isdi⋅sq⋅ =
IsdΨr
Lm–––=
ΨrLr
3
2–pp––– T⋅ =
magCurveInvTorque Ψr_optimal
pr
pL
23Isdmax
fr_optimalRr
2πLr⋅––––– – =

Washing Machine Three-Phase AC Induction Motor Drive, Rev. 0Control Algorithm Overview
Freescale Semiconductor 14Optimal slip control is a common technique used for scalar contro l algorithms where flux and torque
producing components of the stator cannot be contro lled independently. In v ector oriented control
structures we can use a more sophisticated way of optimization based on a motor model using Equation 27 .
This is also the case of our control structure.
3.7 Field-Weakening Control Block
The Field-Weakening Control Block controls the motor magnetizing flux for speeds exceeding the
nominal speed of the motor. The basic task is to ma intain the motor magnetizing flux at a level to prevent
it, exceeding the nominal motor voltage.
Figure 15. Field-Weakening Control
The block has thr ee input quantities:
• actual stator voltage ( voltage_stator )
• actual stator frequency ( f_stator )
• maximum flux producing component of the stator cu rrent evaluated in the loss optimization block
(Isdmax )
Output from the fiel d-weakening block is the required level of the flux producing component of the stator
current ( Isdw ). The internal structure of the field- weakening algorithm is illustrated in Figure 16 .
Figure 16. Internal Structure of Field-Weakening BlockIsdw
IsdmaxField-Weakening
Control
f_stator
voltage_stator
magCurveInvf_stator Ψr_max÷Isdw
IsdmaxLimitervoltage_stator voltage_limit
output saturated

Control Algori thm Overview
Washing Machine Three-Phase AC Induction Motor Drive, Rev. 0
Freescale Semiconductor 15The field-weakening algorithm firs t evaluates a maximum rotor flux ( Ψr_max ) value not exceeding the
voltage limit. It is calculated ac cording to the following formula:
Eqn. 29
Where the DC-Bus voltage is high e nough to generate a mean PWM output voltage higher, or equal to the
motor nominal voltage ( volatge_limit ), the Ustator assigned as equal to voltage_limit . Where
the DC-Bus voltage is low and th e PWM output is saturated, the Ustator is assigned as equal to the
maximum stator voltage ( voltage_stator ), and can be generated by a saturated PWM. The result is
a value setting the maximum rotor flux ( Ψr_max ), and cannot be exceeded. From the magnetizing curve
table, a corresponding direct axis component of the st ator current is evaluated. This axis co mponent is
limited to the Isdmax value. Isdmax is evaluated by the stator winding loss optimization algorithm, setting
the magnetization flux level to minimize power losses in the stator windings.
3.8 Speed Control Loop
The washing machine drum rotational speed is contro lled in a speed control loop. The speed signal is
sensed by means of a tachogenerat or mounted directly on the induc tion motor shaft. The algorithm
evaluates the period of the output tachogenerator voltage signal. Actual speed is evaluated from the signal
period. Actual motor speed is subtra cted from the required speed comma nd, and the regulation error makes
an input to the speed contro ller. The speed controller is implemented as a PID. Output fr om the controller
sets the required value of the motor torque. When the washer drum moves, the wet clot hes inside the drum
bump around, generating high torques rippl es to the motor. To eliminate those ripples and keep the drum
speed as stable as possi ble, a PID controller is used where the derivative com ponents improve the
controller response to the torque ripples . The speed control loop is depicted in Figure 17 .
Figure 17. Speed Control Loopif PWM output is not saturated: Ustator voltage_limit
if PWM output saturated: Ustator voltage_stator
then: Ψr_maxUstator
fstator–––––==
=
-+
PID
actual_speed
actual speed
evaluationspeed_cmd torque_required
tachogenerator comparator

Washing Machine Three-Phase AC Induction Motor Drive, Rev. 0Washing Machine Dri ve Operating Modes
Freescale Semiconductor 163.9 Quadrature Component Evaluation Block
The output of the speed PID controller sets the re quired motor torque. Having the torque command, the
quadrature axis component of the stator current (Isqw) can be evaluated. It is performed in the
“Torque to Isq” block (see Figure 18 ).
Figure 18. Quadrature Component Evaluation Block
Inputs to the control block are the required motor torque ( torque_required ) and required rotor
magnetizing flux space vector amplitude ( Ψrw).
Output from the block is the required va lue of the quadrature axis component (Isqw), serving as an input to
the current control loop.
The quadrature axis component is evaluated considering Equation 9 , therefore defining the relation
between the motor torque (T) and the torque producing component (Isq). To obtain a torque producing
component, Equation 9 is expressed in the following form:
Eqn. 30
The above equation is evaluated within the control block. Together with the direct axis component of the
stator current it defines the required opera ting point of the controlled induction motor.
4 Washing Machine Drive Operating Modes
The washing machine drive runs typically in thr ee different modes of operation. These operating modes
are:
• tumble-wash• out-of-balance detection and load displacement• spin-dry
A typical speed profile of a washi ng machine cycle is illustrated in Figure 19 . The speeds referred to
further in the section relate to a washer drum speed.Torque
to IsqwIsqw
Ψrwtorque_required
isqwTrequired
3
2–ppΨrw––––––- – =

Washing Machine Drive Operating Modes
Washing Machine Three-Phase AC Induction Motor Drive, Rev. 0
Freescale Semiconductor 17Figure 19. Speed Profile of the Washing Cycle
The tumble-wash phase is typical with low drum speed s reversing the direction of the drum rotation every
few turns. Because there are short intervals of rotation, it is necessary to reach a stable rotational speed for
the drum in under two seconds from standstill. This requirement necessi tates a high torque be applied to
the washer drum to make it move. A high generated to rque is one of the key requirements in this operating
mode. The speed of the drum for a tu mble wash is typically 30 – 45 RPM. The exact speed depends on the
type of clothes being washed and is determined by the washing program. The drum speed is low and the
clothes are lifted up within the drum, falling down when they reach the highest point. Wet and heavy
clothes are periodically bumped in the drum, generating high torque ri pples to the motor. The control
algorithm of the drive needs to have enough dynamics to eliminate those ripples. Error in the speed should
not exceed limits of ± 2 RPM. These requirements ca n be satisfied where there is a PID controller for a
speed control loop and an i nner PI current control loop.
The out-of-balance detection and load displacement phase is performed ev ery time before the washer goes
into a spin-dry. The clothes in the drum must be pr operly balanced to minimize centrifugal forces causing
a waggling of the washer. In the first step, the actua l imbalance of the clothes in the washer drum is
detected. The speed of the drum is increased by a ramp up to the va lue at which the clothes become
centrifuged to inner side of the dr um. The algorithm performs an integr ation of the motor torque ripple per
one cycle. The integral value estima tes the size of the load imbalance. If the imbalance is lower than the
safety limit, it starts ramping the speed and goes into a dry-spin. If the imbalance is higher than the safety
limit, the speed of the drum is de creased and the direction of rotation is reversed. The algorithm performs
a new load displacement at the revers ed speed. At the end of a load disp lacement interval the rotation is
reversed, and out-of-balance detection is executed again. The out-of-balance detection and load
displacement sequence is performed until an equal di stribution of the drum load is achieved. Then a
spin-dry is started.
The spin-dry phase is entered if the load imbalance is within safety limits. The drum speed is ramped
steeply until it reaches the require spinning speed. Th e spinning speed differs for particular machines and
washing program. With the presented control it can reach up to 2000 RPM. Once reached, the drum speed 40600
-4010001800RPM
TIME40600
-4010001800RPM
TIME

Washing Machine Three-Phase AC Induction Motor Drive, Rev. 0User Control Interface
Freescale Semiconductor 18is kept constant during the spin-dry interval. When finished, the algorithm pe rforms a non-recuperative
braking. Applying a braking torque, the drum can be stopped faster, thus the washing cycle can be made
shorter. The non-recuperative braking generates a braking torque wi th an energy being dissipated in the
motor windings. It is not lo aded back to the DC-Bus capacitor. No br aking resistor is required in this case
and the hardware design of the power ci rcuit can be significantly simplified.
5 User Control Interface
The washer machine drive demonstra tion is controlled via serial co mmunication protocol (RS232). The
application variables can be monitored in real ti me and drive parameters can be easily modified.
The washer application behavior is controlled through a set of control and status registers and variables.
The master application can set a command for the re quired motor speed and its direction. The status and
control word provides an interface to control the drive operating point. The actual drive status can be
identified as well. It is possible to monitor a wi de range of motor quantitie s on-line. Some of these
quantities are:
• washer drum speed• motor torque• motor voltage and current• DC-Bus voltage• magnetizing flux• direct and quadrature axis comp onents of the stator current
• motor slip frequency• fault status of the application
6 Washer Drive Parameters Tuning
The washer drive application is designed to make the t uning of a particular motor parameters very easy. It
is possible to modify an application for a new motor in a couple of minutes. All the application parameters
are simply accessible through a single parameter file. It is possible to modify all the hardware dependent
constants (current sensing scale, vol tage sensing scale, ove rvoltage and overcurrent limits), application
specific constants (motor speed range, number of tachogenerato r poles, drum-to-motor speed ratio,
out-of-balance detection limits, speed and current c ontroller parameters, etc.), and motor dependent
constants (motor model parameters, number of moto r poles, motor nominal voltage and current, motor
magnetizing curve, motor torque, etc.). All the parameters and constant s are documented for easy
understanding. An example of the conf iguration file showing motor model parameters constants is listed
in Figure 20 .

Freescale Semiconductor Support
Washing Machine Three-Phase AC Induction Motor Drive, Rev. 0
Freescale Semiconductor 19Figure 20. Motor Parame ters Configuration
7 Freescale Semiconductor Support
For more information on the washing machine desi gn, please contact your Freescale representative.
8 References
1. 3-Phase AC Induction Motor Vector C ontrol Using a 56F80x, 56F8100 or 56F8300 Device,
Application Note, AN1930, Freescale Semiconductor Inc. Rev. 2, 2/2005
2. Zeman K., Peroutka Z., Janda M., Automa ticka regulace pohonu s asynchronnimi motory,
University of West Bohemia, Plzen, 2004

Washing Machine Three-Phase AC Induction Motor Drive, Rev. 0Glossary of Symbols
Freescale Semiconductor 209 Glossary of Symbols
α,β – Stator orthogonal coordinate system
d,q – Rotational orthogona l coordinate system
usα,β – Stator voltages in α,β coordinate system
usd,q – Stator voltages in d,q coordinate system
isα,β – Stator currents in α,β coordinate system
isd,q – Stator currents in d,q coordinate system
urα,β – Rotor voltages in α,β coordinate system
urd,q – Rotor voltages in d,q coordinate system
irα,β – Rotor currents in α,β coordinate system
ird,q – Rotor currents in d,q coordinate system
Ψsα,β – Stator magnetic fluxes in α,β coordinate system
Ψsd,q – Stator magnetic fluxes in d,q coordinate system
Ψrα,β – Rotor magnetic fluxes in α,β coordinate system
Ψrd,q – Rotor magnetic fluxes in d,q coordinate system
Rs – Stator phase resistance
Rr – Rotor phase resistance
Ls – Stator phase inductance
Lr – Rotor phase inductance
Lm – Mutual (stator to rotor) inductance
ω / ωs – Electrical rotor angular sp eed / synchronous angular speed
fs – Electrical stator synchronous frequency
fslip – Electrical rotor slip frequency
pp – Number of poles per phase
te – Electromagnetic torque

Glossary of Symbols
Washing Machine Three-Phase AC Induction Motor Drive, Rev. 0
Freescale Semiconductor 21

AN3234
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